Full-bridge and half-bridge compatible driver timing schedule for direct drive backlight system

ABSTRACT

A driver circuit or controller flexibly drives either a half-bridge or a full-bridge switching network in a backlight inverter without modification, redundant circuitry or additional components. The driver circuit includes four outputs to provide four respective driving signals that establish a periodic timing sequence using a zero-voltage switching technique for semiconductor switches in the switching network.

CLAIM FOR PRIORITY

This application is a continuation of U.S. Patent Application Ser. No.11/090,246, filed on Mar. 25, 2005 and entitled “Full-Bridge andHalf-Bridge Compatitle Driver Timing Schedule for Direct Drive BacklightSystem,” which claims the benefit of priority under 35 U.S.C. § 119(e)of U.S. Provisional Application Ser. No. 60/558,512, filed on Apr. 1,2004 and entitled “Full-Bridge and Half-Bridge Compatible Driver TimingSchedule for Direct Drive Backlight System,” each of which is herebyincorporated by reference herein in their entirety.

BACKGROUND

1. Field of the Invention

The invention generally relates to a driver circuit in a backlightsystem for powering fluorescent lamps, and more particularly, relates toa driver circuit with a power efficient timing schedule that canflexibly drive either a half-bridge or a full-bridge switching networkin the backlight system.

2. Description of the Related Art

Fluorescent lamps are used in a number of applications where light isrequired but the power required to generate the light is limited. Oneparticular type of fluorescent lamp is a cold cathode fluorescent lamp(CCFL). CCFLs are used for back or edge lighting of liquid crystaldisplays (LCDs) which are typically found in notebook computers, webbrowsers, automotive and industrial instrumentation, and entertainmentsystems.

A power converter (e.g., an inverter) is typically used to power afluorescent lamp. The inverter includes a controller and a switchingnetwork to convert a direct current (DC) source into an alternatingcurrent (AC) source to power the fluorescent lamp. In a half-bridgeswitching network, a pair of transistors is coupled to the DC source andthe transistors alternately conduct to generate the AC source. In afull-bridge switching network, an arrangement of four transistors iscoupled to the DC source and the transistors conduct in pairs togenerate the AC source. The controller controls transistors in theswitching network. Controllers designed for half-bridge switchingnetworks typically cannot operate full-bridge switching networks, andcontrollers designed for full-bridge switching networks typically do nothave outputs compatible for operating half-bridge networks.

SUMMARY

Embodiments advantageously include driver circuits (or controllers) thatcan switch between half-bridge and full-bridge operations withoutmodification, redundant circuitry or additional components. In oneembodiment, a controller for flexibly driving a half-bridge or afull-bridge switching network in a backlight inverter includes fouroutputs. A first output of the controller provides a first drivingsignal with periodic active and inactive states. A second output of thecontroller provides a second driving signal with active states that arephase shifted by approximately 180° with respect to the active states ofthe first driving signal. The first and the second driving signals havevariable and substantially identical duty cycles that determine relativedurations of the active and the inactive states.

A third output of the controller provides a third driving signal thatsubstantially follows the first driving signal with opposite states andtransition overlaps. For example, the first driving signal and the thirddriving signal are alternately active with overlapping inactive statesduring state transitions. The third driving signal transitions from anactive state to an inactive state before the first driving signaltransitions from an inactive state to an active state. The third drivingsignal also transitions from an inactive state to an active state afterthe first driving signal transitions from an active state to an inactivestate.

A fourth output of the controller provides a fourth driving signal thatsubstantially follows the second driving signal with opposite states andtransitions overlaps. For example, the second driving signal and thefourth driving signal are alternately active with overlapping inactivestates during state transitions. The fourth driving signal transitionsfrom an active state to an inactive state before the second drivingsignal transitions from an inactive state to an active state. The fourthdriving signal also transitions from an inactive state to an activestate after the second driving signal transitions from an active stateto an inactive state.

In one embodiment, a first semiconductor switch (or power transistor)and a second semiconductor switch are arranged in a half-bridgeswitching network of a direct-drive inverter. For example, thesemiconductor switches are coupled between ground and respectiveopposite terminals of a primary winding of a transformer. A power source(e.g., a supply voltage or a current source) is coupled to a center tapof the primary winding of the transformer. A lamp load (e.g., one ormore fluorescent lamps or cold cathode fluorescent lamps) is coupledacross a secondary winding of the transformer.

The semiconductor switches (e.g., N-type transistors) in the half-bridgeswitching network can be advantageously controlled by the first drivingsignal and the second driving signal to generate an AC signal forpowering the lamp load. For example, the first driving signal and thesecond driving signal cause the first semiconductor switch and thesecond semiconductor switch to alternately conduct. Power flows from thepower source to the lamp load in a first polarity when the firstsemiconductor switch is on and the second semiconductor switch is off.Power flows from the power source to the lamp load in a second polaritywhen the second semiconductor switch is on and the first semiconductorswitch is off. Substantially no power flows from the power source to thelamp load when both semiconductor switches are on or off.

In one embodiment, four semiconductor switches are coupled to a primarywinding of a transformer in a full-bridge configuration. The fourdriving signals respectively control the four semiconductor switches togenerate an AC lamp signal for powering a lamp load coupled across asecondary winding of the transformer. For example, the first drivingsignal controls the first semiconductor switch coupled between a firstterminal of the primary winding and ground. The second driving signalcontrols the second semiconductor switch coupled between a secondterminal of the primary winding and ground. The third driving signalcontrols the third semiconductor switch coupled between a power sourceand the first terminal of the primary winding. Finally, the fourthdriving signal controls the fourth semiconductor switch coupled betweenthe power source and the second terminal of the primary winding.

The four driving signals establish a periodic timing sequence thatadvantageously improves power efficiency. For example, the transitionoverlaps between the first and the third driving signals and thetransitions overlaps between the second and the fourth driving signalsfacilitate reduced-voltage (or zero-voltage) switching to improve powerefficiency. Conduction states and idles states are interposed betweenthe different transition overlaps in the periodic timing sequence. Forexample, a first conduction state allows power to flow from the powersource to the lamp load in a first polarity when the first and thefourth semiconductor switches are on while the second and the thirdsemiconductor switches are off. A second conduction state allows powerto flow from the power source to the lamp load in an opposite polaritywhen the first and the fourth semiconductor switches are off while thesecond and the third semiconductor switches are on. Substantially nopower is provided by the power source during the idle states in whichthe first and the second semiconductor switches are on or the third andthe fourth semiconductor switches are on.

In one embodiment, the first and the second semiconductor switches areN-type field-effect-transistors (NFETs) while the third and the fourthsemiconductor switches are P-type FETs (PFETs). Thus, the active statesof the first and the second driving signals correspond to logic highwhile the active states of the third and the fourth driving signalscorrespond to logic low. The third and the fourth driving signals haverising edges that precede respective rising edges of the first and thesecond driving signals by a first duration. The third and the fourthdriving signals have falling edges that trail respective falling edgesof the first and the second driving signals by a second duration.

In one embodiment, the four driving signals are generated from a pair ofinput signals and four delay circuits. For example, a first input signalis provided to a first delay circuit that is coupled in series with asecond delay circuit. A second input signal is provided to a third delaycircuit that is coupled in series with a fourth delay circuit.

In one application in which the first and the second driving signalshave overlapping inactive states, the first delay circuit outputs thefirst driving signal. An output of the second delay circuit is ORed withthe first input signal to generate the third driving signal. The thirddelay circuit outputs the second driving signal. An output of the fourthdelay circuit is ORed with the second input signal to generate thefourth driving signal.

In another application in which the first and the second driving signalshave overlapping inactive states, the first delay circuit outputs thefirst driving signal. The output of the second delay circuit is providedto a first edge-triggered one-shot circuit that has an output coupled toa reset terminal of a first SR latch. The first input signal is providedto a set terminal of the first SR latch. The first SR latch outputs thethird driving signal. The third delay circuit outputs the second drivingsignal. The output of the fourth delay circuit is provided to a secondedge-triggered one-shot circuit that has an output coupled to a resetterminal of a second SR latch. The second input signal is provided to aset terminal of the second SR latch. The second SR latch outputs thefourth driving signal.

In one application in which the first and the second driving signalshave overlapping active states, the output of the first delay circuit isinverted to generate the fourth driving signal. The output of the seconddelay circuit is NORed with the first input signal to generate thesecond driving signal. The output of the third delay circuit is invertedto generate the third driving signal. The output of the fourth delaycircuit is NORed with the second input signal to generate the firstdriving signal.

In another application in which the first and the second driving signalshave overlapping active states, the output of the first delay circuit isinverted to generate the fourth driving signal. The output of the seconddelay circuit is provided to a first one-shot circuit that has an outputcoupled to a reset terminal of a first latch. The first input signal iscoupled to a set terminal of the first latch. The first latch generatesthe second driving signal. The output of the third delay circuit isinverted to generate the third driving signal. The output of the fourthdelay circuit is provided to a second one-shot circuit that has anoutput coupled to a reset terminal of a second latch. The second inputsignal is provided to a set terminal of the second latch. The secondlatch generates the first driving signal.

For purposes of summarizing the invention, certain aspects, advantagesand novel features of the invention have been described herein. It is tobe understood that not necessarily all such advantages may be achievedin accordance with any particular embodiment of the invention. Thus, theinvention may be embodied or carried out in a manner that achieves oroptimizes one advantage or group of advantages as taught herein withoutnecessarily achieving other advantages as may be taught or suggestedherein.

BRIEF DESCRIPTION OF THE DRAWINGS

These drawings and the associated description herein are provided toillustrate embodiments and are not intended to be limiting.

FIG. 1 illustrates one embodiment of a direct drive backlight systemimplemented with a half-bridge switching network.

FIG. 2 illustrates one timing scheme for driving power transistors inthe half-bridge switching network of FIG. 1.

FIG. 3 illustrates one embodiment of a direct drive backlight systemimplemented with a full-bridge switching network.

FIG. 4 illustrates one timing scheme for controlling power transistorsin the full-bridge switching network of FIG. 3.

FIGS. 5(a)-5(h) illustrate one embodiment of a periodic timing sequencefor a full-bridge switching network employing a zero-voltage switchingtechnique to improve power efficiency.

FIG. 6 illustrates one embodiment of driving waveforms to controltransistors in a full-bridge switching network in accordance with theperiodic timing sequence depicted in FIGS. 5(a)-5(h).

FIG. 7 illustrates one embodiment of a controller circuit for generatingthe driving waveforms shown in FIG. 6.

FIG. 8 is a timing diagram for some signals in the controller circuit ofFIG. 7.

FIG. 9 illustrates another embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 6.

FIG. 10 is a timing diagram for some signals in the controller circuitof FIG. 9.

FIGS. 11(a)-11(h) illustrates another embodiment of a periodic timingsequence for a full-bridge switching network that further improves powerefficiency.

FIG. 12 illustrates one embodiment of driving waveforms to controltransistors in a full-bridge switching network in accordance with theperiodic timing sequence depicted in FIGS. 11(a)-11(h).

FIG. 13 illustrates one embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 12.

FIG. 14 illustrates another embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 12.

DETAILED DESCRIPTION OF EMBODIMENTS

Although particular embodiments are described herein, other embodiments,including embodiments that do not provide all of the benefits andfeatures set forth herein, will be apparent to those of ordinary skillin the art.

FIG. 1 illustrates one embodiment of a direct drive backlight systemimplemented with a half-bridge switching network. Two power transistors(or semiconductor switches) 100, 102 are coupled between circuit groundand respective opposite terminals of a primary winding of a transformer104. A power source (VP) is coupled to a center tap of the primarywinding of the transformer 104. The power source can be a supply voltageor a current source. A lamp load 106 is coupled across a secondarywinding of the transformer 104. The lamp load 106 can include one ormore lamps, such as fluorescent lamps or CCFLs. Other half-bridgenetwork configurations including two power transistors are also possibleand may exclude a transformer for coupling to a lamp load.

A controller (not shown) outputs two driving signals to control thesemiconductor switches 100, 102. For example, the first driving signal(Aout) controls the first semiconductor switch (QA) 100 and the seconddriving signal (Bout) controls the second semiconductor switch (QB) 102.The driving signals configured the semiconductor switches 100, 102 toalternately conduct to establish an AC current in the primary windingand the second winding of the transformer 104. In a first conductionstate, power flows from the power source (or supply source) to the lampload 106 in a first polarity when the first semiconductor switch 100 ison and the second semiconductor switch 102 is off. In a secondconduction state, power flows from the power source to the lamp load 106in a second (or opposite) polarity when the second semiconductor switch102 is on and the first semiconductor switch 100 is off. Idle (orpower-off) states can be inserted in between the conduction states.During the idle states, the semiconductor switches 100, 102 are both on(e.g., if the power source is a current source) or both off (e.g., ifthe power source is a voltage source) and substantially no power flowsfrom the power source to the lamp load 106.

FIG. 2 illustrates one timing scheme for driving (or controllingconduction states of) the power transistors 100, 102 in the half-bridgeswitching network of FIG. 1. In the embodiment shown in FIG. 1, thepower transistors 100, 102 are NFETs with driving signals coupled torespective gate terminals of the power transistors 100, 102. Logic highin the driving signals corresponds to turning on the power transistors100, 102 (or an active state) while logic low in the driving signalscorresponds to turning off the power transistors 100, 102 (or aninactive state).

A graph 200 illustrates a first driving signal (Aout) with respect totime for driving the first power transistor 100. A graph 202 illustratesa second driving signal (Bout) with respect to time for driving thesecond power transistor 102. The driving signals are periodically andalternately active (or logic high) for a first predetermined duration(Ta). For example, the first driving signal is active for the firstpredetermined duration during times T1-T2 and T5-T6. The second drivingsignal is active for the first predetermined duration during times T3-T4and T7-T8. Rest periods of a second predetermined duration (Tb) areinserted in between the alternate active states of the driving signals(e.g., during times T2-T3, T4-T5 and T6-T7). The driving signals areboth inactive (or logic low) during the rest periods. Alternately, thedriving signals can be both active during the rest periods.

Thus, the power transistors 100, 102 alternately switch on (or conduct)between periods of rest using the timing scheme illustrated in FIG. 2.Power flows from the power source to the lamp load 106 in a firstpolarity when the first driving signal is active. Power flows from thepower source to the lamp load 106 in a second polarity when the seconddriving signal is active. Substantially no power flows from the powersource to the lamp load 106 when the first and the second drivingsignals are both active or both inactive. The alternating conduction bythe power transistors 100, 102 between the rest periods results in asubstantially AC waveform for powering the lamp load 106. An AC current(or lamp current) flows through a lamp in the lamp load 106 toilluminate the lamp. The brightness or effective power delivered to thelamp is dependent on the power source and switching duty-cycle (i.e.,Ta/Tb).

FIG. 3 illustrates one embodiment of a direct drive backlight systemimplemented with a full-bridge (or H-bridge) switching network. Fourpower transistors 300, 302, 304, 306 are coupled to a primary winding ofa transformer 308. For example, a first pair of power transistors (QA,QB) 300, 302 is coupled between respective opposite terminals of theprimary winding and circuit ground. A second pair of power transistors(QC, QD) 304, 306 is coupled between the respective opposite terminalsof the primary winding and a power source (VP) to complete the H-bridgeswitching network. A lamp load (e.g., a fluorescent lamp) 310 is coupledacross a secondary winding of the transformer 308.

Four driving signals (Aout, Bout, Cout, Dout) respectively control thefour power transistors 300, 302, 304, 306 to generate an AC lamp signalfor powering the lamp load 310 coupled across the secondary winding ofthe transformer 308. For example, the first driving signal (Aout)controls the first power transistors (QA) 300 coupled between a firstterminal of the primary winding and ground. The second driving signal(Bout) controls the second power transistor (QB) 302 coupled between asecond terminal of the primary winding and ground. The third drivingsignal (Cout) controls the third power transistor (QC) 304 coupledbetween the power source and the first terminal of the primary winding.Finally, the fourth driving signal (Dout) controls the fourth powertransistor (QD) 306 coupled between the power source and the secondterminal of the primary winding.

A full-bridge switching network has some advantages over a half-bridgeswitching network. For example, the transformer 308 of FIG. 3 generallycosts less than the transformer 104 of FIG. 1 due to reducedprimary-to-secondary turns ratio and lack of a center tap. Powertransistors used in the full-bridge switching network generally costless than power transistors used in the half-bridge switching networkdue to reduced breakdown voltage requirement. The power transistors inthe half-bridge switching network have a breakdown voltage that iscomparable to at least twice a supply voltage while the powertransistors in the full-bridge switching network have a breakdownvoltage that is comparable to at least the supply voltage.

FIG. 4 illustrates one timing scheme for controlling the powertransistors 300, 302, 304, 306 in the full-bridge switching network ofFIG. 3. In the embodiment shown in FIG. 3, the first pair of powertransistors 300, 302 are NFETs and the second pair of power transistors304, 306 are PFETs. The driving signals (Aout, Bout, Cout, Dout) arecoupled to respective gate terminals of the power transistors 300, 302,304, 306. Logic high in the first two driving signals (Aout, Bout)corresponds to turning on the first pair of power transistors 300, 302(or an active state). Logic low in the last two driving signals (Cout,Dout) corresponds to turning on the second pair of power transistors304, 306 (or an active state).

A graph 400 illustrates the first driving signal (Aout) with respect totime for driving the first power transistor 300. A graph 402 illustratesthe second driving signal (Bout) with respect to time for driving thesecond power transistor 302. A graph 404 illustrates the fourth drivingsignal (Dout) with respect to time for driving the fourth powertransistor 306. A graph 406 illustrates the third driving signal (Cout)with respect to time for driving the third power transistor 304. Thefirst and the second driving signals illustrated in FIG. 4 issubstantially similar to the driving signals illustrated in FIG. 2 forthe half-bridge switching network. The fourth driving signal is aninverted form of the first driving signal, and the third driving signalis an inverted form of the second driving signal. Thus, the first andthe fourth power transistors 300, 306 are switched on and off atapproximately the same times while the second and the third powertransistors 302, 304 are switched on and off at approximately the sametimes.

Referring to FIG. 3, current flows from the second terminal to the firstterminal of the primary winding of the transformer 308 and powertransfers from the power source to the lamp load 310 in a first polarityduring first conduction states when the first driving signal is logichigh (or active) and the fourth driving signal is logic low (or active).Current flows from the first terminal to the second terminal of theprimary winding of the transformer 308 and power transfers from thepower source to the lamp load 310 in a second polarity during secondconduction states when the second driving signal is logic high (oractive) and the third driving signal is logic low (or active).Substantially no power transfers from the power source to the lamp load310 during idle states when the first and the second driving signals areboth inactive (or logic low) as shown in FIG. 4.

FIGS. 5(a)-5(h) illustrate one embodiment of a periodic timing sequencefor the full-bridge switching network of FIG. 3 that employs azero-voltage switching technique to generate an AC lamp signal forpowering the lamp load 310 with improved power efficiency. The powertransistors 300, 302, 304, 306 are represented by schematicallyequivalent single-pole-single-throw switches. The lamp load 310 coupledacross the transformer 308 is not shown for clarity of illustration.

FIG. 5(a) illustrates a first conduction state (or step) in which thefirst power transistor (QA) 300 and the fourth power transistor (QD) 306are on while the second power transistor (QB) 302 and the third powertransistor (QC) 304 are off to allow power to flow from the power source(VP) to the lamp load 310 in a first polarity. For example, currentflows from the power source through the fourth power transistor 306,through the primary winding of the transformer 308 and through the firstpower transistor 300 to ground during the first conduction state. FIGS.5(b)-5(d) illustrate intermediate steps to transition from the firstconduction state to a second conduction state illustrated in FIG. 5(e).

FIG. 5(b) shows a first transition state (or first intermediate step),following the first conduction state, in which the first powertransistor 300 turns off. Because of leakage inductance associated withthe transformer 308, the current through the primary winding of thetransformer 308 does not stop instantaneously. The current flowingthrough the primary winding of the transformer 308 finds a path througha body diode 500 of the third power transistor 304 and back to the powersource. The body diode 500 has an anode coupled to the first terminal ofthe primary winding and a cathode coupled to the power source. With thebody diode 500 conducting, the drain-to-source voltage of the thirdpower transistor 304 is relatively low (e.g., approximately 0.7 volt orone diode voltage drop).

FIG. 5(c) shows a first idle state (or second intermediate step),following the first transition state, in which the third powertransistor 304 turns on. Turning on the third power transistor 304 afterits body diode 500 starts conducting takes advantage of close to zero(or reduced) voltage switching to thereby reduce switching loss. Itshould be noted that although current continues to flow through theprimary winding of the transformer 308 during the idle state, no poweris drawn from the power source.

FIG. 5(d) shows a second transition state (or third intermediate step),following the first idle state, in which the fourth power transistor 306turns off. Similar to the first transition step, the current flowingthrough the primary winding of the transformer 308 does not stopabruptly. The current flowing through the primary winding of thetransformer 308 finds a path from ground through a body diode 502 of thesecond power transistor 302. The body diode 502 has an anode coupled toground and a cathode coupled to the second terminal of the primarywinding.

FIG. 5(e) shows the second conduction state, following the secondtransition state, in which the second power transistor 302 turns on toallow power to flow from the power source to the lamp load 310 in asecond polarity. The second power transistor 302 turns on after its bodydiode 502 starts conducting to take advantage of reduced-voltage (orzero-voltage) switching. In the second conductions state, current flowsfrom the power source through the third power transistor 304, throughthe primary winding of the transformer 308 and through the second powertransistor 302 to ground. The current flows in opposite (or reverse)directions through the primary winding of the transformer 308 betweenthe first and the second conduction states.

FIGS. 5(f)-5(h) illustrate another set of intermediate steps, followingthe same principles shown in FIG. 5(b)-5(d), to transition from thesecond conduction state back to the first conduction state. For example,FIG. 5(f) shows a third transition state, following the secondconduction state, in which the second power transistor 302 turns off andthe current flowing the primary winding of the transformer 308 finds apath to the power source through a body diode 504 of the fourth powertransistor 306. The body diode 504 has an anode coupled to the secondterminal of the primary winding and a cathode coupled to the powersource. FIG. 5(g) shows a second idle state, following the thirdtransition state, in which the fourth power transistor 306 turns onusing zero-voltage switching.

FIG. 5(h) shows a fourth transition state, following the second idlestate, in which the third power transistor 304 turns off and the currentflowing through the primary winding of the transformer 308 finds a pathto ground through a body diode 506 of the first power transistor 300.The body diode 506 has an anode coupled to ground and a cathode coupledto the first terminal of the primary winding. The first power transistor300 turns on using zero-voltage switching in the next step of theperiodic timing sequence to return to the first conduction state. Thezero-voltage switching technique turns on (or closes) a power transistor(or switch) when the voltage across the power transistor (orsource-to-drain voltage of a FET) is at a minimum (or reduced) voltage(e.g., 0.7 volt or substantially zero volt). The zero-voltage switchingtechnique reduces switching power loss due to discharging of thedrain-to-source capacitance associated with turning on the powertransistor.

FIG. 6 illustrates one embodiment of driving waveforms to controltransistors in a full-bridge switching network in accordance with theperiodic timing sequence depicted in FIGS. 5(a)-5(h). For example, acontroller includes four outputs to drive the full-bridge switchingnetwork in a backlight inverter. The controller can also flexibly drivea half-bridge switching network with two of the four outputs. The firstoutput of the controller provides a first driving signal (Aout) withperiodic active and inactive states. The first driving signal has avariable duty-cycle that determines relative durations of the active andthe inactive states, which is one way to control backlight intensity (oramount of power provided to the lamp load 310). A graph 600 illustratesthe first driving signal with respect to time. In one embodiment, thefirst driving signal controls the first power transistor 300 which isshown as an NFET with logic high corresponding to active states. Thegraph 600 shows the first driving signal with periodic active states ofa first duration (Ta) (e.g, from times T1-T2 and T9-T10).

The second output of the controller provides a second driving signal(Bout) that has a substantially identical duty-cycle as the firstdriving signal and is substantially an 180° phase-shifted version of thefirst driving signal. In other words, the active states of the seconddriving signal are phased shifted by approximately 180° with respect tothe active states of the first driving signal to provided complementaryswitching. A graph 602 illustrates the second driving signal withrespect to time. In one embodiment, the second driving signal controlsthe second power transistor 302 which is shown as an NFET with logichigh corresponding to active states. The graph 602 shows the seconddriving signal with periodic active states of the first duration (Ta)(e.g., from times T5-T6 and T13-T14). The active states of the seconddriving signal is phase shifted by 180° from (or occurs in between) theactive states of the first driving signal. The first and the seconddriving signals can advantageously be used to control alternatingconduction by switches in a half-bridge switching network.

The third output of the controller provides a third driving signal(Cout) that substantially follows (or tracks) the first driving signalwith opposite (or opposing) states and transition overlaps. A graph 606shows the third driving signal. In one embodiment, the third drivingsignal controls the third power transistor 304 which is shown as a PFETwith logic low corresponding to active states. With opposing states, thefirst power transistor 300 and the third power transistor 304 arealternately on. With transition overlaps, the third power transistor 304turns off before the first power transistor 300 turns on and the thirdpower transistor 304 turns on after the first power transistor 300 turnsoff.

The graph 606 shows the third driving signal with periodic inactivestates that exceed the first duration (e.g., from times T0-T3 andT8-T11). Thus, the third driving signal is substantially similar to thefirst driving signal except the leading (or rising) edge of the thirddriving signal precedes the leading edge of the first driving signal bya first overlapping duration and the trailing (or falling) edge of thethird driving signal succeeds the trailing edge of the first drivingsignal after a second overlapping duration. In other words, the thirddriving signal transitions from an active state (i.e., logic low) to aninactive state (i.e., logic high) before the first driving signaltransitions from an inactive state. (i.e., logic low) to an active state(i.e., logic high). The third driving signal also transitions from aninactive state to an active state after the first driving signaltransitions from an active state to an inactive state. During the firstand the second overlapping durations, the first and the third drivingsignals are both in inactive states.

The fourth output of the controller provides a fourth driving signal(Dout) that substantially follows the second driving signal withopposite states and transition overlaps. A graph 604 shows the fourthdriving signal. In one embodiment, the fourth driving signal controlsthe fourth power transistor 306 which is shown as a PFET with logic lowcorresponding to active states. With opposite states, the second powertransistor 302 and the fourth power transistor 306 are alternately on.With transition overlaps, the fourth power transistor 306 turns offbefore the second power transistor 302 turns on and the fourth powertransistor 306 turns on after the second power transistor 302 turns off.

The graph 604 shows the fourth driving signal with periodic inactivestates that exceed the first duration (e.g., from times T4-T7 andT12-T15). Thus, the fourth driving signal is substantially similar tothe second driving signal except the leading edge of the fourth drivingsignal precedes the leading edge of the second driving signal by a thirdoverlapping duration and the trailing edge of the fourth driving signalsucceeds the trailing edge of the second driving signal after a fourthoverlapping duration. In other words, the fourth driving signaltransitions from an active state (i.e., logic low) to an inactive state(i.e., logic high) before the second driving signal transitions from aninactive state (i.e., logic low) to an active state (i.e., logic high).The fourth driving signal also transitions from an inactive state to anactive state after the second driving signal transitions from an activestate to an inactive state. During the third and the fourth overlappingdurations, the second and the fourth driving signals are both ininactive states. FIG. 6 shows the four overlapping durations to havesubstantially identical time lengths (i.e., To). However, each of theoverlapping durations can be a different time length.

Referring to FIG. 6 in conjunction with FIGS. 5(a)-5(h), the period ofoverlapping active states between the first and the fourth drivingsignals (e.g., from time T1-T2 or T9-T10) corresponds to the firstconduction state shown in FIG. 5(a). The trailing edge transitionoverlaps between the first and the third driving signals (e.g., fromtimes T2-T3 and T10-T11) correspond to the first transition state shownin FIG. 5(b). The first period of overlapping inactive states (or firstrest period) between the first and the second driving signals (e.g.,from time T3-T4 or T11-T12) corresponds to the first idle state shown inFIG. 5(c). The leading edge transition overlaps between the second andthe fourth driving signals (e.g., from times T4-T5 and T12-T13)correspond to the second transition state shown in FIG. 5(d). The periodof overlapping active states between the second and the third drivingsignals (e.g., from time T5-T6 or T13-T14) corresponds to the secondconduction state shown in FIG. 5(e). The trailing edge transitionoverlaps between the second and the fourth driving signals (e.g., fromtimes T6-T7 and T14-T15) correspond to the third transition state shownin FIG. 5(f). The second period of overlapping inactive states (orsecond rest period) between the first and the second driving signals(e.g., from time T7-T8) corresponds to the second idle state shown inFIG. 5(g). Finally, the leading edge transition overlaps between thefirst and the third driving signals (e.g., from times T0-T1 and T8-T9)correspond to the fourth transition state shown in FIG. 5(h).

As discussed above, power is drawn from the power source and deliveredto the lamp load 310 through the transformer 308 during the first andthe second conduction states (or power-on states). No net current flowsout of the power source during the first and the second idle states (orpower-off states). In addition to facilitating power efficiency byreduced-voltage switching, the four transition states help avoidshoot-through current associated with the first power transistor 300 andthe third power transistor 304 (or the second power transistor 302 andthe fourth power transistor 306) being on at substantially the sametime. The duration of the transition states (or transition overlaps) arechosen to guarantee that one of the power transistors is turned offbefore the other power transistor is turned on.

FIG. 7 illustrates one embodiment of a controller circuit for generatingthe driving waveforms shown in FIG. 6. The controller circuit of FIG. 7accepts two input signals (A, B) with overlapping logic low levels (orinactive states) and generates four driving signals (Aout, Bout, Cout,Dout). For example, the two input signals are substantially similar tothe driving signals shown in FIG. 2 for driving a half-bridge switchingnetwork. The first and the second driving signals (Aout, Bout) also haveoverlapping logic low levels (or inactive states).

In one embodiment, a first delay circuit 700 and a second delay circuit702 are coupled in series to the first input signal (A) to generate thefirst driving signal (Aout) and the third driving signal (Cout). Forexample, the first delay circuit 700 receives the first input signal anddelays the first input signal by a first time delay (To(1)) to generatethe first driving signal. The second delay circuit 702 receives thefirst driving signal and adds a second time delay (To(2)) to generate afirst twice-delayed signal (A_delay). The first twice-delayed signal andthe first input signal are provided to a first logic OR circuit (orgate) 708 to generate the third driving signal.

In a similar configuration, a third delay circuit 704 and a fourth delaycircuit 706 are coupled in series to the second input signal (B) togenerate the second driving signal (Bout) and the fourth driving signal(Dout). For example, the third delay circuit 704 receives the secondinput signal and delays the second input signal by a third time delay(To(3)) to generate the second driving signal. The fourth delay circuit706 receives the second driving signal and adds a fourth time delay(To(4)) to generate a second twice-delayed signal (B_delay). The secondtwice-delayed signal and the second input signal are provided to asecond logic OR circuit 710 to generate the fourth driving signal. Thetime delays for the respective delay circuits 700, 702, 704, 706 can besubstantially identical or different.

FIG. 8 is a timing diagram for some signals in the controller circuit ofFIG. 7. A graph 800 shows the first input signal (A) with respect totime. A graph 802 shows the first driving signal (Aout) with respect totime. A graph 804 shows the first twice-delayed signal (A_delay) withrespect to time. Finally, a graph 806 shows the third driving signal(Cout) with respect to time.

The first input signal has periodic active states or periods of logichigh levels (e.g., from times T0-T3 and T6-T9). The first driving signalsubstantially follows the first input signal with leading and trailingedge transitions delayed by the first time delay (To(1)). The firsttwice-delayed signal substantially follows the first driving signal withleading and trailing edge transitions further delayed by the second timedelay (To(2)). The third driving signal has leading edge transitionsfollow the leading edge transitions of the first input signal andtrailing edge transitions follow the trailing edge transitions of thefirst twice-delayed signal. Thus, the third driving signal has leadingedge transitions that precede the leading edge transitions of the firstdriving signal by the first time delay and trailing edge transitionsthat succeed the trailing edge transitions of the first driving signalby the second time delay.

One possible disadvantage of the controller circuit shown in FIG. 7 islimited duty cycle for the driving signals. The pulse width of the inputsignals cannot be shorter than any of the time delays. In other words,duration of conduction states (e.g., logic high periods for the firstdriving signal) cannot be shorter than duration of transition states(e.g., delay in edge transitions between the first and the third drivingsignals or time delays of the delay circuits 700, 702, 704, 706).

FIG. 9 illustrates another embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 6. The circuitimplementation of FIG. 9 advantageously allows the duration of theconduction states to be shorter than the durations of the transitionstates. A first delay circuit 900 and a second delay circuit 902 arecoupled in series to a first input signal (A) to generate a firstdriving signal (Aout) and a third driving signal (Cout). For example,the first delay circuit 900 receives the first input signal and adds afirst time delay (To(1)) to generate the first driving signal. Thesecond delay circuit 902 receives an output of the first delay circuit900 and adds a second time delay (To(2)) to generate a firsttwice-delayed signal (A_delay). The first twice-delayed signal isprovided to a first one-shot circuit (e.g., a falling edge-triggeredmonostable circuit) 908. An output of the first one-short circuit 908 isprovided to a reset terminal of a first SR latch 912. The first inputsignal is provided to a set terminal of the first SR latch 912. Thefirst SR latch 912 outputs the third driving signal (e.g., at its Qoutput).

In a similar configuration, a third delay circuit 904 and a fourth delaycircuit 906 are coupled in series to a second input signal (B) togenerate a second driving signal (Bout) and a fourth driving signal(Dout). For example, the third delay circuit 904 receives the secondinput signal and adds a third time delay (To(3)) to generate the seconddriving signal. The fourth delay circuit 906 receives an output of thethird delay circuit 904 and adds a fourth time delay (To(4)) to generatea second twice-delayed signal (B_delay). The second twice-delayed signalis provided to a second one-shot circuit 910. An output of the secondone-shot circuit 910 is provided to a reset terminal of a second SRlatch 914. The second input signal is provided to a set terminal of thesecond SR latch 914. The second SR latch 914 outputs the fourth drivingsignal.

FIG. 10 is a timing diagram for some signals in the controller circuitof FIG. 9. A graph 1000 shows the first input signal (A) with respect totime. A graph 1002 shows the first driving signal (Aout) with respect totime. A graph 1004 shows the first twice-delayed signal with respect totime. Finally, a graph 1006 shows the third driving signal (Cout) withrespect to time.

The first input signal has periodic durations of logic high levels(e.g., from times T0-T1 and T6-T7). The first driving signalsubstantially follows the first input signal with rising and fallingedge transitions delayed by the first time delay (To(l)). The firsttwice-delayed signal substantially follows the first driving signal withrising and falling edge transitions further delayed by the second timedelay (To(2)). In the timing diagrams shown in FIG. 10, the logic highduration of the first input signal is less than the duration of thefirst time delay or the second time delay. The rising edge of the firstinput signal sets the rising edge of the third driving signal and thefirst SR latch 912 holds the logic high level of the third drivingsignal until the falling edge of the first twice-delayed signal resetsthe first SR latch 912 using the first one-shot circuit 908. Thus,similar to the circuit implementation of FIG. 7, the third drivingsignal has rising edge transitions that precede the rising edgetransitions of the first driving signal by the first time delay andfalling edge transitions that succeed the falling edge transitions ofthe first driving signal by the second time delay. However, unlike thecircuit implementation of FIG. 7, the circuit implementation of FIG. 9does not have a duty cycle limitation.

FIGS. 11(a)-11(h) illustrate another embodiment of a periodic timingsequence for a full-bridge switching network that further improves powerefficiency. FIGS. 11(a)-11(h) are substantially similar to FIGS.5(a)-5(h) with exception of the idle states shown in FIGS. 5(c) and5(g). As described above, no net current flows out of the power sourceduring the idle (or power-off) states. However, current is flowingthrough the primary winding of the transformer 308 and power continuesto be delivered to the lamp load 310. The power delivered to the lampload 310 during the power-off states comes from energy stored in theleakage inductance of the transformer 308. During the power-off states,power efficiency is limited by the on-resistance of conductingtransistors. The conducting transistors in FIGS. 5(c) and 5(g) are thethird and the fourth power transistors 304, 306, which are PFETs. It isoften easier and cheaper to find NFETs with lower on-resistance thanPFETs.

FIGS. 11(a)-11(h) shows the periodic timing sequence in which the firstand the second power transistors (e.g., NFETs) 300, 302 are on duringthe power-off states to further improve power efficiency. For example,FIG. 11(a) illustrates a first conduction state in which the firsttransistor (QA) 300 and the fourth power transistor (QD) 306 are onwhile the second transistor (QB) 302 and the third power transistor (QC)304 are off to allow power to flow from the power source (VP) to thelamp load 310 in a first polarity. For example, current flows from thepower source through the fourth power transistor 306, through theprimary winding of the transformer 308 and through the first powertransistor 300 to ground during the first conduction state. FIGS.11(b)-11(d) illustrate intermediate steps to transition from the firstconduction state to a second conduction state illustrated in FIG. 11(e).

FIG. 11(b) shows a first transition state, following the firstconduction state, in which the fourth power transistor 306 turns off.Because of leakage inductance associated with the transformer 308, thecurrent through the primary winding of the transformer 308 does not stopinstantaneously. The current flowing through the primary winding of thetransformer 308 finds a path to ground through a body diode 502 of thesecond power transistor 302. The body diode 502 has a cathode coupled tothe second terminal of the primary winding and an anode coupled toground. With the body diode 502 conducting, the source-to-drain voltageof the second power transistor 302 is relatively low (e.g.,approximately 0.7 volt or one diode voltage drop).

FIG. 11(c) shows a first idle state, following the first transitionstate, in which the second power transistor 302 turns on. FIG. 11(d)shows a second transition state, following the first idle state, inwhich the first power transistor 300 turns off. Similar to the firsttransition step, the current flowing through the primary winding of thetransformer 308 does not stop abruptly. The current flowing through theprimary winding of the transformer 308 finds a path through a body diode500 of the third power transistor 304 back to the power source. The bodydiode 500 has a cathode coupled to the power source and an anode coupledto the first terminal of the primary winding.

FIG. 11(e) shows the second conduction state, following the secondtransition state, in which the third power transistor 304 turns on toallow power to flow from the power source to the lamp load 310 in asecond polarity. The third power transistor 302 turns on after its bodydiode 500 starts conducting to take advantage of reduced-voltageswitching. In the second conductions state, current flows from the powersource through the third power transistor 304, through the primarywinding of the transformer 308 and through the second power transistor302 to ground. The current flows in opposite directions through theprimary winding of the transformer 308 between the first and the secondconduction states.

FIGS. 11(f)-11(h) illustrate another set of intermediate steps,following the same principles shown in FIG. 11(b)-11(d), to transitionfrom the second conduction state back to the first conduction state. Forexample, FIG. 11(f) shows a third transition state, following the secondconduction state, in which the third power transistor 304 turns off andthe current flowing the primary winding of the transformer 308 finds apath to ground through a body diode 506 of the first power transistor300. The body diode 506 has a cathode coupled to the first terminal ofthe primary winding and an anode coupled to ground. FIG. 11(g) shows asecond idle state, following the third transition state, in which thefirst power transistor 300 turns on using zero-voltage switching. Thus,NFETs with relatively lower on-resistance are conducting during thefirst and the second idle states.

FIG. 11(h) shows a fourth transition state, following the second idlestate, in which the second power transistor 302 turns off and thecurrent flowing through the primary winding of the transformer 308 findsa path to the power source through a body diode 504 of the fourth powertransistor 306. The body diode 504 has a cathode coupled to the powersource and an anode coupled to the second terminal of the primarywinding. The fourth power transistor 306 turns on using zero-voltageswitching in the next step of the periodic timing sequence to return tothe first conduction state.

FIG. 12 illustrates one embodiment of driving waveforms to controltransistors in a full-bridge switching network in accordance with theperiodic timing sequence depicted in FIGS. 11(a)-11(h). For example, acontroller outputs four driving signals to flexibly drive either ahalf-bridge or a full-bridge switching network using a reduced-voltage(or zero-voltage) switching technique. A graph 1200 shows a firstdriving signal (Aout) with respect to time. A graph 1202 shows a seconddriving signal (Bout) with respect to time. A graph 1204 shows a fourthdriving signal (Dout) with respect to time. Finally a graph 1206 shows athird driving signal (Cout) with respect to time.

The driving signals shown in FIG. 12 are substantially similar to thedriving signals shown in FIG. 6 except the first and the second drivingsignals have overlapping active states (e.g., from times T3-T4, T7-T8and T11-T12) while the third and the fourth driving signals haveoverlapping inactive states to allow the first and the second powertransistors (NFETs) 300, 302 to conduct during the idle states. Thefirst and the second driving signals have substantially identical activeand inactive durations phase-shifted by approximately 180°. The thirdand the first driving signals have tracking logic levels (or oppositestates) and transition overlaps. That is, the leading edges of the thirddriving signal precedes the respective leading edges of the firstdriving signal by a first overlap duration (e.g., from time T6-T7 orT14-T15) and the trailing edges of the third driving signal succeeds therespective trailing edges of the first driving signal by a secondoverlap duration (e.g., from time T4-T5 or T12-T13). The second and thefourth driving signals also have tracking logic levels and transitionoverlaps. That is, the leading edges of the fourth driving signalprecedes the respective leading edges of the second driving signal by athird overlap duration (e.g., from time T2-T3 or T10-T11) and thetrailing edges of the fourth driving signal succeeds the respectivetrailing edges of the second driving signal by a fourth overlap duration(e.g., from time T0-T1 or T8-T9).

FIG. 13 illustrates one embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 12. The controllercircuit of FIG. 13 accepts two input signals (A, B) with overlappinglogic low levels and generates four driving signals (Aout, Bout, Cout,Dout). In one embodiment, the two input signals are substantiallysimilar to driving signals for driving a half-bridge switching network.The first and the second driving signals (Aout, Bout) have overlappinglogic high levels (or active states) in the controller circuit of FIG.13.

In one embodiment, a first delay circuit 1300 and a second delay circuit1302 are coupled in series to the first input signal (A) to generate thesecond driving signal (Bout) and the fourth driving signal (Dout). Forexample, the first delay circuit 1300 receives the first input signaland delays the first input signal by a first time delay. A firstinverter 1308 is coupled to an output of the first delay circuit 1300 togenerate the fourth driving signal. The second delay circuit 1302 iscoupled to the output of first delay circuit 1300 and adds a second timedelay to generate a first twice-delayed signal. The first twice-delayedsignal and the first input signal are provided to a first logic NORcircuit (or gate) 1310 to generate the second driving signal.

In a similar configuration, a third delay circuit 1304 and a fourthdelay circuit 1306 are coupled in series to the second input signal (B)to generate the first driving signal (Aout) and the third driving signal(Cout). For example, the third delay circuit 1304 receives the secondinput signal and delays the second input signal by a third time delay. Asecond inverter 1312 is coupled to an output of the third delay circuit1304 to generate the third driving signal. The fourth delay circuit 1306is coupled to the output of the third delay circuit 1304 and adds afourth time delay to generate a second twice-delayed signal. The secondtwice-delayed signal and the second input signal are provided to asecond logic NOR circuit 1314 to generate the first driving signal. Thetime delays for the respective delay circuits 1300, 1302, 1304, 1306 canbe substantially identical (e.g., To) or different.

FIG. 14 illustrates another embodiment of a controller circuit forgenerating the driving waveforms shown in FIG. 12. A first delay circuit1400 and a second delay circuit 1402 are coupled in series to a firstinput signal (A) to generate a second driving signal (Bout) and a fourthdriving signal (Dout). For example, the first delay circuit 1400receives the first input signal and adds a first time delay. A firstinverter is coupled to an output of the first delay circuit 1400 togenerate the fourth driving signal. The second delay circuit 1402receives the output of the first delay circuit 1400 and adds a secondtime delay to generate a first twice-delayed signal. The firsttwice-delayed signal is provided to a first one-shot circuit 1410. Anoutput of the first one-short circuit 1410 is provided to a resetterminal of a first latch 1412. The first input signal is provided to aset terminal of the first latch 1412. The first latch 1412 outputs thesecond driving signal (e.g., at its QB output).

In a similar configuration, a third delay circuit 1404 and a fourthdelay circuit 1406 are coupled in series to a second input signal (B) togenerate a first driving signal (Aout) and a third driving signal(Cout). For example, the third delay circuit 1404 receives the secondinput signal and adds a third time delay. A second inverter 1414 iscoupled to an output of the third delay circuit 1404 to generate thethird driving signal. The fourth delay circuit 1406 receives the outputof the third delay circuit 1404 and adds a fourth time delay to generatea second twice-delayed signal. The second twice-delayed signal isprovided to a second one-shot circuit 1416. An output of the secondone-shot circuit 1416 is provided to a reset terminal of a second latch1418. The second input signal is provided to a set terminal of thesecond latch 1418. The second latch 1418 outputs the first drivingsignal. The circuit implementation of FIG. 14 advantageously has nolimitation on the duty cycle of the driving signals.

Various embodiments have been described above. Although described withreference to these specific embodiments, the descriptions are intendedto be illustrative and are not intended to be limiting. Variousmodifications and applications may occur to those skilled in the artwithout departing from the true spirit and scope of the invention asdefined by the appended claims.

1. An inverter comprising: a switching network comprising a plurality ofsemiconductor switches that alternately conduct to convert a directcurrent source into an alternating current source to power a load; and acontroller that outputs a plurality of driving signals to control thesemiconductor switches, wherein a first driving signal and a thirddriving signal are alternately active with overlapping inactive statesduring state transitions, a second driving signal and a fourth drivingsignal are alternately active with overlapping inactive states duringstate transitions, and the active states of the first driving signal andthe second driving signal are phase shifted by approximately 180°. 2.The inverter of claim 1, further comprising a transformer, wherein thesemiconductor switches are coupled to a primary winding of thetransformer and the load is coupled to a secondary winding of thetransformer.
 3. The inverter of claim 1, wherein the load comprises atleast one cold cathode fluorescent lamp configured for backlighting aliquid crystal display.
 4. The inverter of claim 1, wherein theswitching network is a half-bridge switching network comprising twosemiconductor switches and the first driving signal and the seconddriving signal are used to control the two semiconductor switches. 5.The inverter of claim 1, wherein the switching network is a full-bridgeswitching network comprising four semiconductor switches, the firstdriving signal and the fourth driving signal are used to control the twosemiconductor switches that complete a first conduction path to deliverpower to the load in a first polarity, and the second driving signal andthe third driving signal are used to control the two semiconductorswitches that complete a second conduction path to deliver power to theload in a second polarity:
 6. The inverter of claim 1, wherein thecontroller comprises four delay circuits to generate the drivingsignals.
 7. A method to flexibly control a half-bridge or a full-bridgeswitching network in a backlight inverter, the method comprisinggenerating at least four driving signals with alternating states tocontrol the switching network, wherein the first and the second drivingsignals are substantially identical with an 180° phase shift, the thirdand the fourth driving signals have rising edges that precede respectiverising edges of the first and the second driving signals by a firstduration, and the third and the fourth driving signals have fallingedges that trail respective falling edges of the first and the seconddriving signals by a second duration.
 8. The method of claim 7, whereinthe driving signals are generated using a pair of input signals and fourdelay circuits.
 9. The method of claim 7, wherein the half-bridgenetwork comprises two semiconductor switches and the first and thesecond driving signals control the two semiconductor switches.
 10. Themethod of claim 7, wherein the full-bridge network comprises foursemiconductor switches, the first and the fourth driving signals controlthe two semiconductor switches that complete a first conduction path todeliver power to a load, and the second and the third driving signalscontrol the two semiconductor switches that complete a second conductionpath to deliver power to the load.